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FeaTures
n
LT3956 80VIN, 80VOUT Constant-Current, Constant-Voltage Converter DescripTion
The LT(R)3956 is a DC/DC converter designed to operate as a constant-current source and constant-voltage regulator. It is ideally suited for driving high current LEDs. It features an internal low side N-channel power MOSFET rated for 84V at 3.3A and driven from an internal regulated 7.15V supply. The fixed frequency, current-mode architecture results in stable operation over a wide range of supply and output voltages. A ground referenced voltage FB pin serves as the input for several LED protection features, and also makes it possible for the converter to operate as a constant-voltage source. A frequency adjust pin allows the user to program the frequency from 100kHz to 1MHz to optimize efficiency, performance or external component size. The LT3956 senses output current at the high side of the LED string. High side current sensing is the most flexible scheme for driving LEDs, allowing boost, buck mode or buck-boost mode configuration. The PWM input provides LED dimming ratios of up to 3000:1, and the CTRL input provides additional analog dimming capability.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 7199560 and 7321203.
3000:1 True Color PWMTM Dimming n Wide Input Voltage Range: 4.5V to 80V n Output Voltage Up to 80V n Internal 3.3A/84V Switch n Constant-Current and Constant-Voltage Regulation n 250mV High Side Current Sense n Drives LEDs in Boost, Buck Mode, Buck-Boost Mode, SEPIC or Flyback Topology n Adjustable Frequency: 100kHz to 1MHz n Open LED Protection n Programmable Undervoltage Lockout with Hysteresis n Constant-Voltage Loop Status Pin n PWM Disconnect Switch Driver n CTRL Pin Adjusts High Side Current Sense Threshold n Low Shutdown Current: <1A n Programmable Soft-Start n Available in the 36-Lead (5mm x 6mm) QFN Package
applicaTions
n n n
High Power LED Battery Charger Accurate Current Limited Voltage Regulator
Typical applicaTion
94% Efficient 25W White LED Headlamp Driver
VIN, 6V TO 60V (80V TRANSIENT) 2.2F 2 332k 100k 332k INTVCC 100k VMODE PWM SS RT VC 28.7k 375kHz 34k 4.7nF 40.2k VIN EN/UVLO VREF LT3956 CTRL ISN 1M FB 16.2k PWMOUT GND INTVCC INTVCC 4.7F M1
3956 TA01a
Efficiency vs VIN
100
22H SW PGND ISP
D1 2.2F 5
96
0.68 370mA
EFFICIENCY (%)
25W LED STRING
92
88
84 80
0
20
47nF
40 VIN (V)
60
80
3956 TA01b
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LT3956 absoluTe MaxiMuM raTings
(Note 1)
pin conFiguraTion
TOP VIEW VMODE CTRL PWM VREF
VIN, ISP ISN ..............................................................80V , SW ............................................................................84V EN/UVLO (Note 3) .....................................................80V INTVCC ...................................................... VIN + 0.3V, 8V PWMOUT ..................................................INTVCC + 0.3V CTRL, PWM, VMODE ................................................12V FB ...............................................................................8V VC, VREF , SS ................................................................3V RT ............................................................................1.5V PGND to GND.........................................................0.5V Operating Junction Temperature Range (Note 2) ............................................. -40C to 125C Maximum Junction Temperature........................... 125C Storage Temperature Range................... -65C to 125C Lead Temperature (Soldering, 10 sec) .................. 300C
SS
RT
36 35 34 33 32 31 30 NC 1 EN/UVLO 2 INTVCC 3 GND 4 VIN 6 SW 8 SW 9 NC 10 12 13 14 15 16 17 PGND PGND PGND PGND PGND PGND 38 SW 37 GND 28 ISP 27 ISN 25 FB 24 GND 23 PWMOUT 21 SW 20 SW
UHE PACKAGE 36-LEAD (5mm 6mm) PLASTIC QFN TJMAX = 125C, JA = 43C/W, JC = 5C/W EXPOSED PAD (PIN 37) IS GND, MUST BE SOLDERED TO PCB EXPOSED PAD (PIN 38) IS SW, MUST BE SOLDERED TO PCB
orDer inForMaTion
LEAD FREE FINISH LT3956EUHE#PBF LT3956IUHE#PBF TAPE AND REEL LT3956EUHE#TRPBF LT3956IUHE#TRPBF PART MARKING* 3956 3956 PACKAGE DESCRIPTION 36-Lead (5mm x 6mm) Plastic QFN 36-Lead (5mm x 6mm) Plastic QFN TEMPERATURE RANGE -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ This product is only offered in trays. For more information go to: http://www.linear.com/packaging/
elecTrical characTerisTics
PARAMETER VIN Minimum Operating Voltage VIN Shutdown IQ VIN Operating IQ (Not Switching) VREF Voltage VREF Line Regulation
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
CONDITIONS VIN Tied to INTVCC EN/UVLO = 0V EN/UVLO = 1.15V PWM = 0V -100A IVREF 0A 4.5V VIN 80V
l l
MIN
VC
TYP 0.1 1.4
MAX 4.5 1 5 1.7 2.045
UNITS V A A mA V %/V
1.965
2.00 0.006
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LT3956 elecTrical characTerisTics
PARAMETER SW Pin Leakage SW Pin Current Limit SW Pin Voltage Drop SS Pull-Up Current Error Amplifier Full-Scale Current Sense Threshold ( V(ISP-ISN)) Current Sense Threshold at CTRL = 1V ( V(ISP-ISN)) Current Sense Threshold at CTRL = 0.5V ( V(ISP-ISN)) Current Sense Threshold at CTRL = 0.1V ( V(ISP-ISN)) CTRL Range for Current Sense Threshold Adjustment CTRL Input Bias Current Current Sense Amplifier Input Common Mode Range ( VISN) ISP/ISN Short-Circuit Threshold ( V(ISP-ISN)) ISP/ISN Short-Circuit Fault Sensing Common Mode Range ( VISN) ISP/ISN Input Bias Current (Combined) LED Current Sense Amplifier gm VC Output Impedance VC Standby Input Bias Current FB Regulation Voltage (VFB) FB Amplifier gm FB Pin Input Bias Current FB Voltage Loop Active Threshold FB Overvoltage Threshold Oscillator Switching Frequency SW Minimum Off-Time SW Minimum On-Time Linear Regulator INTVCC Regulation Voltage Dropout (VIN - INTVCC) INTVCC Undervoltage Lockout INTVCC Current Limit INTVCC Current in Shutdown EN/UVLO = 0V, INTVCC = 7V IINTVCC = -10mA, VIN = 7V
l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
CONDITIONS SW = 48V
l
MIN 3.3 8
l l l l
TYP 5 3.9 220 10 250 225 100 0 50
MAX 10 4.6 13 257 231 103 4.5 1.1 100 80
UNITS A A mV A mV mV mV mV V nA V mV V A A S k
I(SW) = 2A Current Out of Pin FB = 0V, ISP = 48V, CTRL 1.2V CTRL = 1V, FB = 0V, ISP = 48V CTRL = 0.5V CTRL = 0.1V, FB = 0V, ISP = 48V Current Out of Pin, CTRL = 0V 2.9 ISN = 0V 300 0 PWM = 5V (Active), ISP = ISN = 48V PWM = 0V (Standby), ISP = ISN = 48V 1V < VC < 2V PWM = 0V ISP = ISN = 0V, 48V FB = VFB, ISP = ISN Current Out of Pin, FB = 1V VMODE Falling PWMOUT Falling VFB - 65mV VFB + 50mV
l
240 217 96 -2.5 0
335
370 3
80 0 120 15000 -20 1.220 1.232 1.250 1.250 480 40 VFB - 50mV VFB + 60mV 100 1000 170 200 7 7.15 1 4.1 14 17 8
0.1
20 1.270 1.265 100 VFB - 40mV VFB + 80mV 125 1050
nA V V S nA V V
RT = 100k RT = 10k
l
90 925
kHz kHz ns ns
7.3 4.4 25 12
V V V mA A
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LT3956 elecTrical characTerisTics
PARAMETER Logic Inputs/Outputs PWM Threshold Voltage PWM Pin Resistance to GND EN/UVLO Threshold Voltage Falling EN/UVLO Rising Hysteresis EN/UVLO Input Low Voltage EN/UVLO Pin Bias Current Low EN/UVLO Pin Bias Current High VMODE Output Low (VOL) VMODE Pin Leakage PWMOUT Driver tr PWMOUT Driver Output Rise Time tf PWMOUT Driver Output Fall Time PWMOUT Output Low (VOL) PWMOUT Output High (VOH) Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3956E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C INTVCC - 0.05 CL = 560pF CL = 560pF 35 35 0.05 ns ns V V IVIN Drops Below 1A EN/UVLO = 1.15V EN/UVLO = 1.30V IVMODE = 1mA FB = 0V, VMODE = 12V 0.1 1.7 2.1 10
l l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
CONDITIONS MIN 0.85 45 1.185 TYP 1.35 60 1.220 20 0.4 2.5 100 200 5 1.245 MAX 1.8 UNITS V k V mV V A nA mV A
to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3956I is guaranteed to meet performance specifications over the -40C to 125C operating junction temperature range. Note 3: For VIN below 6V, the EN/UVLO pin must not exceed VIN for proper operation.
Typical perForMance characTerisTics
V(ISP-ISN) Threshold vs VCTRL
300 250 V(ISP - ISN) THRESHOLD (mV) V(ISP - ISN) THRESHOLD (mV) 200 150 100 50 0 -50 0 0.5 1 1.5 CTRL VOLTAGE (V) 2
3956 G01
TA = 25C, unless otherwise noted. V(ISP-ISN) Full-Scale Threshold vs Temperature
256 254 V(ISP -ISN) THRESHOLD (mV) 252 250 248 246 244 242 -50 50 25 0 75 TEMPERATURE (C) 125
3956 G03
V(ISP-ISN) Threshold vs VISP with Reduced CTRL Voltage
103 102 101 100 99 98 97 CTRL = 0.5V
CTRL = 2V
0
20
40 60 ISP VOLTAGE (V)
80
3956 G02
-25
100
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LT3956 Typical perForMance characTerisTics
FB Regulation Voltage (VFB) vs Temperature
1.28 1.27 1.26 VREF (V) 1.25 VFB (V) 1.24 1.23 1.22 1.21 1.20 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125
3956 G04
TA = 25C, unless otherwise noted.
VREF Voltage vs Temperature
2.04 2.03 2.02 2.01 2.00 1.99 1.98 1.97 1.96 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125
3956 G05
VREF Voltage vs VIN
2.04 2.03 2.02 2.01 VREF (V) 2.00 1.99 1.98 1.97 1.96 0 20 40 VIN (V) 60 80
3956 G06
Switching Frequency vs RT
10000 500
Switching Frequency vs Temperature
RT = 26.7k 2.4
EN/UVLO Hysteresis Current vs Temperature
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
450 IEN/UVLO (A) 125
3956 G08
2.2
1000
400
2.0
100
350
1.8
10
10 RT (k)
100
3956 G07
300 -50
-25
50 25 0 75 TEMPERATURE (C)
100
1.6 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3956 G09
Quiescent Current vs VIN
2.0 PWM = 0V 4.4
SW Pin Current Limit vs Temperature
1.28 1.26 EN/UVLO VOLTAGE (V)
EN/UVLO Threshold vs Temperature
VIN CURRENT (mA)
CURRENT LIMIT (A)
1.5
4.2
EN/UVLO RISING 1.24 1.22 1.20 1.18 -50 EN/UVLO FALLING
1.0
4.0
0.5
3.8
0
0
20
40 VIN (V)
60
80
3956 G10
3.6 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3956 G11
-25
50 25 0 75 TEMPERATURE (C)
100
125
3956 G12
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LT3956 Typical perForMance characTerisTics
Quiescent Current vs Switching Frequency
12 10 VIN CURRENT (mA) 8 6 4 2 0 20
TA = 25C, unless otherwise noted.
INTVCC Current Limit vs Temperature
NOT SWITCHING 7.4
INTVCC Voltage vs Temperature
INTVCC CURRENT LIMIT (mA)
18
7.3 INTVCC (V) -25 50 25 0 75 TEMPERATURE (C) 100 125
16
7.2
14
12 10 -50
7.1
0
800 600 400 200 SWITCHING FREQUENCY (kHz)
1000
3956 G13
7.0 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3956 G14
3956 G15
SW Pin Current Limit vs Duty Cycle
4.5 312.5
LED Current Sense Threshold vs FB Voltage
VCTRL = 2V 0 -0.5 LDO DROPOUT (V)
INTVCC Dropout Voltage vs INTVCC Current
V(ISP-ISN) THRESHOLD (mV)
SW PIN CURRENT LIMIT (A)
4.0
250.0
-40C 25C
187.5
-1.0
3.5
125.0
-1.5
125C
3.0
62.50
-2.0 -2.5
2.5
0
25
50 75 DUTY CYCLE (%)
100
3956 G16
0 1.2
1.22
1.24 1.26 FB VOLTAGE (V)
1.28
3956 G17
0
3
6 9 LDO CURRENT (mA)
12
15
3956 G18
ISP/ISN Input Bias Current vs CTRL Voltage
80 ISP INPUT BIAS CURRENT (A) ON-RESISTANCE (m) 60 180 160 140 120 100 80 60 40 20 0 0 0.5 1 CTRL (V) 1.5 2
3956 G19
Switch On-Resistance vs Temperature
PWM INPUT PWMOUT 5V/DIV
PWMOUT Waveform
CPWMOUT = 2.2nF
40 ISN 20
200ns/DIV -25 0 25 50 75 100 125
3956 G21
0 -50
TEMPERATURE (C)
3956 G20
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LT3956 pin FuncTions
NC: No Internal Connection. These pins may be left floating or connected to an adjacent pin. EN/UVLO: Shutdown and Undervoltage Detect Pin. An accurate 1.22V falling threshold with externally programmable hysteresis detects when power is OK to enable switching. Rising hysteresis is generated by the external resistor divider and an accurate internal 2.1A pull-down current. Above the 1.24V (nominal) threshold (but below 6V), EN/UVLO input bias current is sub-A. Below the falling threshold, a 2.1A pull-down current is enabled so the user can define the hysteresis with the external resistor selection. An undervoltage condition resets soft-start. Tie to 0.4V, or less, to disable the device and reduce VIN quiescent current below 1A. INTVCC: Regulated supply for internal loads, GATE driver and PWMOUT driver. Supplied from VIN and regulates to 7.15V (typical). INTVCC must be bypassed with a 4.7F capacitor placed close to the pin. Connect INTVCC directly to VIN if VIN is always less than or equal to 7V. GND: Ground. The exposed pad, Pin 37, is ground and must be soldered directly to the ground plane. VIN: Input Supply Pin. Must be locally bypassed with a 0.22F (or larger) capacitor to PGND placed close to the IC. SW: The exposed pad, Pin 38, is the drain of the switching N-channel MOSFET and must be connected to the external inductor. PGND: Source terminal of switch and the GND input to the switch current comparator. Kelvin connect to the GND plane close to the IC using Pin 12. Pins 13 to 17 should be connected externally to the PGND terminals of components in the switching path. See the Board Layout section. PWMOUT: Buffered Version of the PWM Signal. This pin is used to drive the LED load disconnect N-channel MOSFET or level shift. This pin also serves in a protection function for the FB overvoltage condition--will toggle if the FB input is greater than the FB regulation voltage (VFB) plus 60mV (typical). The PWMOUT pin is driven from INTVCC. Use of a MOSFET with gate cut-off voltage higher than 1V is recommended. FB: Voltage Loop Feedback Pin. FB is intended for constant-voltage regulation or for LED protection/open LED detection. The internal transconductance amplifier with output VC will regulate FB to 1.25V (nominal) through the DC/DC converter. If the FB input is regulating the loop, the VMODE pull-down is asserted. This action may signal an open LED fault. If FB is driven above the FB threshold (by an external power supply spike, for example), the VMODE pull-down will be de-asserted and the PWMOUT pin will be driven low to protect the LEDs from an overcurrent event. Do not leave the FB pin open. If not used, connect to GND. ISN: Connection point for the negative terminal of the current feedback resistor. If ISN is greater than 2.9V, the LED current can be programmed by ILED = 250mV/RLED when VCTRL > 1.2V or ILED = (VCTRL -100mV)/(4 * RLED) if VCTRL < 1V. Input bias current is typically 20A. Below 3V, ISN is an input to the short-circuit protection feature that forces GATE to 0V if ISP exceeds ISN by more than 350mV (typ). ISP: Connection point for the positive terminal of the current feedback resistor. Input bias current for this pin depends on CTRL pin voltage, as shown in the Typical Performance Characteristics. ISP is an input to the short-circuit protection feature when ISN is less than 3V. VC: Transconductance Error Amplifier Output Pin. This pin is used to stabilize the voltage loop with an RC network. This pin is high impedance when PWM is low, a feature that stores the demand current state variable for the next PWM high transition. Connect a capacitor between this pin and GND; a resistor in series with the capacitor is recommended for fast transient response. CTRL: Current Sense Threshold Adjustment Pin. Regulating threshold V(ISP - ISN) is 0.25 * VCTRL plus an offset for 0V < VCTRL < 1V. For VCTRL > 1.2V the current sense threshold is constant at the full-scale value of 250mV. For 1V < VCTRL < 1.2V, the dependence of the current sense threshold upon VCTRL transitions from a linear function to a constant value, reaching 98% of full-scale value by VCTRL = 1.1V. Connect CTRL to VREF for the 250mV default threshold. Do not leave this pin open.
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LT3956 pin FuncTions
VREF: Voltage Reference Output Pin (typically 2V). This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of LED load. Can supply up to 100A. PWM: A signal low turns off switcher, idles oscillator and disconnects VC pin from all internal loads. PWMOUT pin follows PWM pin. PWM has an internal pull-down resistor. If not used, connect to INTVCC. VMODE: An open-collector pull-down on VMODE asserts if the FB input is greater than the FB regulation threshold minus 50mV (typical). To function, the pin requires an external pull-up resistor. When the PWM input is low and the DC/DC converter is idle, the VMODE condition is latched to the last valid state when the PWM input was high. When PWM input goes high again, the VMODE pin will be updated. This pin may be used to report an open LED fault. Use a pull-up current less than 1mA. SS: Soft-Start Pin. This pin modulates oscillator frequency and compensation pin voltage (VC) clamp. The soft-start interval is set with an external capacitor. The pin has a 10A (typical) pull-up current source to an internal 2.5V rail. The soft-start pin is reset to GND by an undervoltage condition (detected by EN/UVLO pin) or thermal limit. RT: Switching Frequency Adjustment Pin. Set the frequency using a resistor to GND (for resistor values, see the Typical Performance curve or Table 1). Do not leave the RT pin open.
block DiagraM
+ -
EN/UVLO 1.22V 2.1A A6 SHDN 1.31V FB VC PWMOUT PWM LDO 7.15V INTVCC VIN
1.25V
1.25V SHORT-CIRCUIT DETECT
+ gm -
SCILMB
A5
10A AT FB = 1.25V SCILMB
+ -
ISN ISP CTRL
350mV
+ A10 -
5k
10A
1.1V
+ + A3 -
CTRL BUFFER Q2
+ A1 -
gm EAMP
10A AT A1+ = A1-
+ A2 -
R S
PWM COMPARATOR
VC SSCLAMP
RAMP GENERATOR 100kHz TO 1MHz OSCILLATOR
20k FAULT LOGIC TSD 165C 10A
VREF
1.25V
1.2V FREQ PROG FB
2V
A7
170k
3956 BD
SS
RT
+ -
140A
+ + -
+ -
OVFB COMPARATOR
+ -
A8
SW
Q
DRIVER ISENSE
+ A4 -
PGND
GND
VMODE 1mA (MAX)
+ -
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LT3956 operaTion
The LT3956 is a constant-frequency, current mode converter with a low side N-channel MOSFET switch. The switch and PWMOUT pin drivers, and other chip loads, are powered from INTVCC, which is an internally regulated supply. In the discussion that follows, it will be helpful to refer to the Block Diagram of the IC. In normal operation, with the PWM pin low, the power switch is turned off and the PWMOUT pin is driven to GND, the VC pin is high impedance to store the previous switching state on the external compensation capacitor, and the ISP and ISN pin bias currents are reduced to leakage levels. When the PWM pin transitions high, the PWMOUT pin transitions high after a short delay. At the same time, the internal oscillator wakes up and generates a pulse to set the PWM latch, turning on the internal power MOSFET switch. A voltage input proportional to the switch current, sensed by an internal current sense resistor, is added to a stabilizing slope compensation ramp and the resulting switch-current sense signal is fed into the positive terminal of the PWM comparator. The current in the external inductor increases steadily during the time the switch is on. When the switch-current sense voltage exceeds the output of the error amplifier, labeled VC , the latch is reset and the switch is turned off. During the switch off phase, the inductor current decreases. At the completion of each oscillator cycle, internal signals such as slope compensation return to their starting points and a new cycle begins with the set pulse from the oscillator. Through this repetitive action, the PWM control algorithm establishes a switch duty cycle to regulate a current or voltage in the load. The VC signal is integrated over many switching cycles and is an amplified version of the difference between the LED current sense voltage, measured between ISP and ISN, and the target difference voltage set by the CTRL pin. In this manner, the error amplifier sets the correct peak switch-current level to keep the LED current in regulation. If the error amplifier output increases, more current is demanded in the switch; if it decreases, less current is demanded. The switch current is monitored during the on-phase and is not allowed to exceed the current limit threshold of 3.9A (typical). If the SW pin exceeds the current limit threshold, the SR latch is reset regardless of the output state of the PWM comparator. Likewise, at an ISP/ISN common mode voltage less than 3V, the difference between ISP and ISN is monitored to determine if the output is in a short-circuit condition. If the difference between ISP and ISN is greater than 335mV (typical), the SR latch will be reset regardless of the PWM comparator. These functions are intended to protect the power switch, as well as various external components in the power path of the DC/DC converter. In voltage feedback mode, the operation is similar to that described above, except the voltage at the VC pin is set by the amplified difference of the internal reference of 1.25V (nominal) and the FB pin. If FB is lower than the reference voltage, the switch current will increase; if FB is higher than the reference voltage, the switch demand current will decrease. The LED current sense feedback interacts with the FB voltage feedback so that FB will not exceed the internal reference and the voltage between ISP and ISN will not exceed the threshold set by the CTRL pin. For accurate current or voltage regulation, it is necessary to be sure that under normal operating conditions, the appropriate loop is dominant. To deactivate the voltage loop entirely, FB can be connected to GND. To deactivate the LED current loop entirely, the ISP and ISN should be tied together and the CTRL input tied to VREF . Two LED specific functions featured on the LT3956 are controlled by the voltage feedback pin. First, when the FB pin exceeds a voltage 50mV lower (-4%) than the FB regulation voltage, the pull-down driver on the VMODE pin is activated. This function provides a status indicator that the load may be disconnected and the constant-voltage feedback loop is taking control of the switching regulator. When the FB pin exceeds the FB regulation voltage by 60mV (5% typical), the PWMOUT pin is driven low, ignoring the state of the PWM input. In the case where the PWMOUT pin drives a disconnect NFET, this action isolates the LED load from GND, preventing excessive current from damaging the LEDs. If the FB input exceeds the overvoltage threshold (1.31V typical), then an externally driven overvoltage event may have caused the FB pin to be too high and the VMODE pull-down will be deactivated until the FB pin drops below the overvoltage threshold.
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LT3956 applicaTions inForMaTion
INTVCC Regulator Bypassing and Operation The INTVCC pin requires a capacitor for stable operation and to store the charge for the switch driver and PWMOUT pin switching currents. Choose a 10V rated low ESR, X7R or X5R ceramic capacitor for best performance. A 4.7F capacitor will be adequate for many applications. Place the capacitor close to the IC to minimize the trace length to the INTVCC pin and also to the IC ground. An internal current limit on the INTVCC output protects the LT3956 from excessive on-chip power dissipation. The INTVCC pin has its own undervoltage disable (UVLO) set to 4.1V (typical) to protect the internal MOSFET from excessive power dissipation caused by not being fully enhanced. If the INTVCC pin drops below the UVLO threshold, the PWMOUT pin will be forced to 0V, the power switch turned off and the soft-start pin will be reset. If the input voltage, VIN, will not exceed 7V, then the INTVCC pin could be connected to the input supply. This action allows the LT3956 to operate from as low as 4.5V. Be aware that a small current (less than 12A) will load the INTVCC in shutdown. Otherwise, the minimum operating VIN value is determined by the dropout voltage of the linear regulator and the 4.4V (4.1V typical) INTVCC undervoltage lockout threshold mentioned above. Programming the Turn-On and Turn-Off Thresholds With the EN/UVLO Pin The falling UVLO value can be accurately set by the resistor divider. A small 2.1A pull-down current is active when EN/UVLO is below the falling threshold. The purpose of this current is to allow the user to program the rising hysteresis. The following equations should be used to determine the values of the resistors: VIN,FALLING = 1.22 * R1 + R2 R2 LED Current Programming The LED current is programmed by placing an appropriate value current sense resistor, RLED , between the ISP and ISN pins. Typically, sensing of the current should be done at the top of the LED string. If this option is not available, then the current may be sensed at the bottom of the string, but take caution that the minimum ISN value does not fall below 3V, which is the lower limit of the LED current regulation function. The CTRL pin should be tied to a voltage higher than 1.2V to get the full-scale 250mV (typical) threshold across the sense resistor. The CTRL pin can also be used to dim the LED current to zero, although relative accuracy decreases with the decreasing voltage sense threshold. When the CTRL pin voltage is less than 1V, the LED current is: ILED = VCTRL - 100mV RLED * 4
When the CTRL pin voltage is between 1V and 1.2V the LED current varies with CTRL, but departs from the previous equation by an increasing amount as the CTRL voltage increases. Ultimately, above CTRL = 1.2V, the LED current no longer varies with CTRL. At CTRL = 1.1V, the actual value of ILED is ~98% of the equation's estimate. When VCTRL is higher than 1.2V, the LED current is regulated to: ILED = 250mV RLED
VIN,RISING = 2.1A * R1 + VIN,FALLING
VIN LT3956 EN/UVLO R2
3956 F01
R1
Figure 1
The CTRL pin should not be left open (tie to VREF if not used). The CTRL pin can also be used in conjunction with a thermistor to provide overtemperature protection for the LED load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high LED load current, low switching frequency and/or a smaller value output filter capacitor. Some level of ripple signal is acceptable: the compensation capacitor on the VC pin filters the signal so the average difference between ISP and ISN is regulated to the user-programmed value. Ripple voltage amplitude (peak-to-peak) in excess of
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0
LT3956 applicaTions inForMaTion
20mV should not cause misoperation, but may lead to noticeable offset between the average value and the userprogrammed value. Output Current Capability An important consideration when using a switch with a fixed current limit is whether the regulator will be able to supply the load at the extremes of input and output voltage range. Several equations are provided to help determine this capability. Some margin to data sheet limits is included. For boost converters: IOUT(MAX ) 2.5A VIN(MIN) VOUT(MAX ) Programming Output Voltage (Constant-Voltage Regulation) or Open LED/Overvoltage Threshold For a boost or SEPIC application, the output voltage can be set by selecting the values of R3 and R4 (see Figure 2) according to the following equation: VOUT = 1.25 * R3 + R4 R4
For a boost type LED driver, set the resistor from the output to the FB pin such that the expected voltage level during normal operation will not exceed 1.1V. For an LED driver of buck mode or a buck-boost mode configuration, the output voltage is typically level-shifted to a signal with respect to GND as illustrated in Figure 3. The output can be expressed as: VOUT = VBE + 1.25 * R3 R4
+
VOUT LT3956 FB R4
3956 F03
For buck mode converters: IOUT(MAX) 2.5A For SEPIC and buck-boost mode converters: VIN(MIN) IOUT(MAX ) 2.5A ( VOUT(MAX ) + VIN(MIN) ) These equations assume the inductor value and switching frequency have been selected so that inductor ripple current is ~600mA. Ripple current higher than this value will reduce available output current. Be aware that current limited operation at high duty cycle can greatly increase inductor ripple current, so additional margin may be required at high duty cycle. If some level of analog dimming is acceptable at minimum supply levels, then the CTRL pin can be used with a resistor divider to VIN (as shown on page 1) to provide a higher output current at nominal VIN levels.
VOUT LT3956 FB R4
3956 F02
R3
RLED LED ARRAY
COUT
-
100k
Figure 3. Feedback Resistor Connection for Buck Mode or Buck-Boost Mode LED Driver
ISP/ISN Short-Circuit Protection Feature for SEPIC The ISP and ISN pins have a protection feature independent of the LED current sense feature that operates at ISN below 3V. The purpose of this feature is to provide continuous current sensing when ISN is below the LED current sense common mode range (during start-up or an output short-circuit fault) to prevent the development of excessive switching currents that could damage the power components in a SEPIC converter. The action threshold (335mV, typ) is above the default LED current sense threshold, so that no interference will occur over the ISN voltage range where these two functions overlap. This feature acts in the same manner as switch-current limit -- it prevents switch turn-on until the ISP/ISN difference falls below the threshold.
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R3
Figure 2. Feedback Resistor Connection for Boost or SEPIC LED Drivers
LT3956 applicaTions inForMaTion
Dimming Control There are two methods to control the current source for dimming using the LT3956. One method uses the CTRL pin to adjust the current regulated in the LEDs. A second method uses the PWM pin to modulate the current source between zero and full current to achieve a precisely programmed average current. To make this method of current control more accurate, the switch demand current is stored on the VC node during the quiescent phase when PWM is low. This feature minimizes recovery time when the PWM signal goes high. To further improve the recovery time, a disconnect switch may be used in the LED current path to prevent the ISP node from discharging during the PWM signal low phase. The minimum PWM on or off time will depend on the choice of operating frequency through the RT input. For best overall performance, the minimum PWM low or high time should be at least six switching cycles (6s for fSW = 1MHz). Programming the Switching Frequency The RT frequency adjust pin allows the user to program the switching frequency from 100kHz to 1MHz to optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. For an appropriate RT resistor value see Table 1. An external resistor from the RT pin to GND is required--do not leave this pin open.
Table 1. Switching Frequency vs RT Value
fOSC (kHz) 1000 900 800 700 600 500 400 300 200 100 RT (k) 10 11.8 13 15.4 17.8 21 26.7 35.7 53.6 100
300 250 200 TIME (ns) 150 100 50 0 -50 MINIMUM ON-TIME MINIMUM OFF-TIME
Duty Cycle Considerations Switching duty cycle is a key variable defining converter operation, therefore, its limits must be considered when programming the switching frequency for a particular application. The fixed minimum on-time and minimum off-time (see Figure 4) and the switching frequency define the minimum and maximum duty cycle of the switch, respectively. The following equations express the minimum/maximum duty cycle: Min Duty Cycle = (minimum on-time) * switching frequency Max Duty Cycle = 1 - (minimum off-time) * switching frequency When calculating the operating limits, the typical values for on/off-time in the data sheet should be increased by at least 60ns to allow margin for PWM control latitude and SW node rise/fall times.
-25
50 25 0 75 TEMPERATURE (C)
100
125
3956 F04
Figure 4. Typical Switch Minimum On and Off Pulse Width vs Temperature
Thermal Considerations The LT3956 is rated to a maximum input voltage of 80V. Careful attention must be paid to the internal power dissipation of the IC at higher input voltages to ensure that a junction temperature of 125C is not exceeded. This junction limit is especially important when operating at high ambient temperatures. If the LT3956's junction temperature reaches 165C (typ), the power switch will be turned off and the soft-start (SS) pin will be discharged to GND. Switching
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LT3956 applicaTions inForMaTion
will be enabled after the device temperature drops 10C. This function is intended to protect the device during momentary overload conditions. The major contributors to internal power dissipation are the current in the linear regulator to drive the switch, and the ohmic losses in the switch. The linear regulator power is proportional to VIN and switching frequency, so at high VIN the switching frequency should be chosen carefully to ensure that the IC does not exceed a safe junction temperature. The internal junction temperature of the IC can be estimated by: TJ = TA + [VIN * (IQ + fSW * 7nC) + ISW2 * 0.14 * DSW] * JA where TA is the ambient temperature, IQ is the quiescent current of the part (maximum 1.7mA) and JA is the package thermal impedance (43C/W for the 5mm x 6mm QFN package). For example, an application with TA(MAX) = 85C, VIN(MAX) = 60V, fSW = 400kHz, and having an average switching current of 2.5A at 70% duty cycle, the maximum IC junction temperature will be approximately: TJ = 85C + [(2.5A)2 * 0.14 * 0.7 + 60V * (1.7mA + 400kHz * 7nC)] * 43C/W= 123C The Exposed Pads on the bottom of the package must be soldered to a plane. These should then be connected to internal copper planes with thermal vias placed directly under the package to spread out the heat dissipated by the IC. Open LED Detection The LT3956 provides an open-drain status pin, VMODE, that pulls low when the FB pin is within ~50mV of its 1.25V regulated voltage. If the open LED clamp voltage is programmed correctly using the FB pin, then the FB pin should never exceed 1.1V when LEDs are connected, therefore, the only way for the FB pin to be within 50mV of the regulation voltage is for an open LED event to have occurred. Input Capacitor Selection The input capacitor supplies the transient input current for the power inductor of the converter and must be placed and sized according to the transient current requirements. The switching frequency, output current and tolerable input voltage ripple are key inputs to estimating the capacitor value. An X7R type ceramic capacitor is usually the best choice since it has the least variation with temperature and DC bias. Typically, boost and SEPIC converters require a lower value capacitor than a buck mode converter. Assuming that a 100mV input voltage ripple is acceptable, the required capacitor value for a boost converter can be estimated as follows: V 1F CIN(F ) = ILED( A ) * OUT * TSW(s) * VIN A * s Therefore, a 4.7F capacitor is an appropriate selection for a 400kHz boost regulator with 12V input, 48V output and 1A load. With the same VIN voltage ripple of 100mV, the input capacitor for a buck converter can be estimated as follows: CIN(F ) = ILED( A ) * TSW (s) * 4 . 7 F A * s
A 10F input capacitor is an appropriate selection for a 400kHz buck mode converter with a 1A load. In the buck mode configuration, the input capacitor has large pulsed currents due to the current returned through the Schottky diode when the switch is off. In this buck converter case it is important to place the capacitor as close as possible to the Schottky diode and to the PGND return of the switch. It is also important to consider the ripple current rating of the capacitor. For best reliability, this capacitor should have low ESR and ESL and have an adequate ripple current rating. The RMS input current for a buck mode LED driver is: IIN(RMS) = ILED *
( 1 - D) * D
where D is the switch duty cycle.
Table 2. Recommended Ceramic Capacitor Manufacturers
MANUFACTURER TDK Kemet Murata Taiyo Yuden WEB SITE www.tdk.com www.kemet.com www.murata.com www.t-yuden.com
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LT3956 applicaTions inForMaTion
Output Capacitor Selection The selection of the output capacitor depends on the load and converter configuration, i.e., step-up or step-down and the operating frequency. For LED applications, the equivalent resistance of the LED is typically low and the output filter capacitor should be sized to attenuate the current ripple. Use of an X7R type ceramic capacitor is recommended. To achieve the same LED ripple current, the required filter capacitor is larger in the boost and buck-boost mode applications than that in the buck mode applications. Lower operating frequencies will require proportionately higher capacitor values. Soft-Start Capacitor Selection For many applications, it is important to minimize the inrush current at start-up. The built-in soft-start circuit significantly reduces the start-up current spike and output voltage overshoot. The soft-start interval is set by the softstart capacitor selection according to the equation: TSS = CSS * 2V 10A it is important to consider diode leakage, which increases with the temperature, from the output during the PWM low interval. Therefore, choose the Schottky diode with sufficiently low leakage current. Table 3 has some recommended component vendors.
Table 3. Schottky Rectifier Manufacturers
VENDOR On Semiconductor Diodes, Inc. Central Semiconductor WEB SITE www.onsemi.com www.diodes.com www.centralsemi.com
Inductor Selection The inductor used with the LT3956 should have a saturation current rating appropriate to the maximum switch current of 4.6A. Choose an inductor value based on operating frequency, input and output voltage to provide a current mode signal of approximately 0.6A magnitude. The following equations are useful to estimate the inductor value (TSW = 1/fOSC): LBUCK = TSW * VLED VIN - VLED VIN * 0.6 A
(
)
A typical value for the soft-start capacitor is 0.01F The . soft-start pin reduces the oscillator frequency and the maximum current in the switch. The soft-start capacitor is discharged when EN/UVLO falls below its threshold, during an overtemperature event or during an INTVCC undervoltage event. During start-up with EN/UVLO, charging of the soft-start capacitor is enabled after the first PWM high period. Schottky Rectifier Selection The power Schottky diode conducts current during the interval when the switch is turned off. Select a diode rated for the maximum SW voltage of the application and the RMS diode current. If using the PWM feature for dimming,
LBUCK-BOOST = LBOOST =
(
TSW * VLED * VIN VLED + VIN * 0.6 A
)
TSW * VIN VLED - VIN VLED * 0.6 A
(
)
Table 4 provides some recommended inductor vendors.
Table 4. Inductor Manufacturers
VENDOR Sumida Wurth Elektronik Coiltronics Renco Coilcraft WEB SITE www.sumida.com www.we-online.com www.cooperet.com www.rencousa.com www.coilcraft.com
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LT3956 applicaTions inForMaTion
Loop Compensation The LT3956 uses an internal transconductance error amplifier whose VC output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability. The inductor and output capacitor are chosen based on performance, size and cost. The compensation resistor and capacitor at VC are selected to optimize control loop response and stability. For typical LED applications, a 4.7nF compensation capacitor at VC is adequate, and a series resistor should always be used to increase the slew rate on the VC pin to maintain tighter regulation of LED current during fast transients on the input supply to the converter. Board Layout The high speed operation of the LT3956 demands careful attention to board layout and component placement. The exposed pads of the package are important for thermal management of the IC. It is crucial to achieve a good electrical and thermal contact between the GND exposed pad and the ground plane of the board. To reduce electromagnetic
CSS RT 36 35 34 33 32 31 30 1 2 CVCC VIAS TO GND PLANE VIAS TO SW PLANE VIN R1 R2 3 4 GND 6 8 9 10 12 13 14 15 16 17 PGND VIAS L1 COUT SW 25 24 23 21 20 R4 R3 1 LT3956 28 27 VIA FROM LED+ LED- 3 M1 2 VIAS FROM PGND
interference (EMI), it is important to minimize the area of the high dV/dt switching node between the inductor, SW pin and anode of the Schottky rectifier. Use a ground plane under the switching node to eliminate interplane coupling to sensitive signals. The lengths of the high dI/dt traces: 1) from the switch node through the switch to PGND, and 2) from the switch node through the Schottky rectifier and filter capacitor to PGND, should be minimized. The ground points of these two switching current traces should come to a common point then connect to the ground plane at the PGND pin of the LT3956 through a separate via to Pin 12, as shown in the suggested layout (Figure 5). Likewise, the ground terminal of the bypass capacitor for the INTVCC regulator should be placed near the GND of the IC. The ground for the compensation network and other DC control signals should be star connected to the GND Exposed Pad of the IC. Do not extensively route high impedance signals such as FB and VC, as they may pick up switching noise. Since there is a small variable DC input bias current to the ISN and ISP inputs, resistance in series with these pins should be minimized to avoid creating an offset in the current sense threshold.
VMODE PWM CTRL CC RC VIA FROM VOUT
D1
COUT
RS CVIN VIN PGND VOUT LED+ VIA VIA
LED+
3956 F05
Figure 5. Boost Converter Suggested Layout
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LT3956 Typical applicaTions
94% Efficient 25W White LED Headlamp Driver
VIN 6V TO 60V (80V TRANSIENT) CVIN 2.2F 2 R1 332k R2 100k VIN EN/UVLO 332k 40.2k VREF LT3956 CTRL INTVCC 100k VMODE PWM SS RT VC RC 20k CC 4.7nF ISN FB R3 1M R4 16.2k PWMOUT GND INTVCC INTVCC CVCC 4.7F 25W LED STRING (CURRENT DERATED FOR VIN < 11V) L1 22H SW PGND ISP RS 0.68 370mA
D1 COUT 2.2F 5 M1: VISHAY SILICONIX Si2328DS D1: DIODES INC PDS5100 L1: COILTRONICS DR125-220 C1, C2: MURATA GRM42-2x7R225
RT 28.7k 375kHz CSS 47nF
M1
3956 TA02a
SEE SUGGESTED LAYOUT (FIGURE 5)
PWM Waveforms for 25W Headlamp Driver
PWM
ILED 200mA/DIV
ILI 1A/DIV VOUT = 68V VIN = 15V 5s/DIV
3956 TA02b
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LT3956 Typical applicaTions
Buck-Boost Mode LED Driver
VIN 9V TO 45V L1 68H C1 4.7F 1M VIN EN/UVLO 187k VREF LT3956 INTVCC 100k CTRL ISN 619k FB VMODE PGND PWM SS RT PWMOUT VC GND INTVCC 3.4k 10nF INTVCC 4.7F Q1 1k
3956 TA03a
Efficiency vs VIN
VOUT 4.7F 35V VIN EFFICIENCY (%) 680m 100 96
D1 SW ISP 1F 100V
92
10k
24V LED STRING 350mA
88
84 80
0
10
35.7k 300kHz 0.1F
750 M1 VIN
20 30 VIN (V)
40
50
3956 TA03b
L1: COILCRAFT MSS1038-683 D1: ON SEMICONDUCTOR MBRS3100T3 M1: ZETEX ZXM6IP03F Q1: ZETEX FMMT493
28VIN /0V to 28V SEPIC SuperCap Charger with Input Current Limit
VIN 28V 1.2A 1F VIN EN/UVLO LT3956 PWMOUT VMODE 10k INTVCC PWM VC C2 4.7F
3956 TA04a
200m
L1A 33H C1 10F 1:1
C4 10F
Input and Output Current vs Output Voltage
VOUT 0V TO 28V 3.0 2.5 2.0 1.5 1.0 INPUT 0.5 0 OUTPUT
D1
ISP
ISN SW PGND FB
C3 10F
536k 25k
1M 40.2k INTVCC 2k Q1 59k 1M
CTRL VREF SS RT GND 28.7k 375kHz 14k
INPUT/OUTPUT CURRENT (A)
L1B
0
5
10
15 20 VOUT (V)
25
30
3956 TA04b
10nF
30.1k L1: WURTH ELEKTRONIK 744871330 D1: ON SEMI MBRS36OT Q1: MMBTA42 C1, C3, C4: TAIYO-YUDEN GMK 3I6BJ106
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LT3956 Typical applicaTions
90% Efficient, 20W SEPIC LED Driver
L1A 33H C1 4.7F 50V 1:1 VIN EN/UVLO 185k VREF CTRL INTVCC 100k VMODE PWM SS RT VC 28.7k 375kHz 0.01F 15k 10nF 25k LT3956 ISN 1M FB PWMOUT GND INTVCC C2 4.7F 10V 56.2k 20W LED STRING CURRENT DERATED FOR VIN < 13V M1
3956 TA05a
Efficiency vs VIN
D1 C3 10F 2 35V 100 96 EFFICIENCY (%)
VIN 8V TO 50V
C4 2.2F (50V)
1M
250k
SW PGND ISP
L1B
92
88
0.25
1A
84 80
0
10
20 30 VIN (V)
40
50
3956 TA05b
L1: COILTRONICS DRQ127-330 D1: VISHAY PDS5100 M1: ZETEX ZXM61N03F
90W Buck Mode LED Driver, 80VIN /60VOUT
VIN 64V TO 80V 100 1M EN/UVLO INTVCC 100k VMODE VREF 24.3k CTRL 13k PWM SS RT VC 1k 20k LT3956 ISN FB M1 PWMOUT Q2 10k 16 WHITE LEDs, 90W L1 33H SW PGND GND INTVCC INTVCC 4.7F D1 VIN C1 2.2F 4
3956 TA06a
Efficiency vs VIN
VIN
ISP 470 0.1 1.5A Q1
267k
200k 200k
C2 2.2F 3 EFFICIENCY (%)
98
96
94
92 90
64
68
72 VIN (V)
76
80
3956 TA05b
28.7k 375kHz 0.1F
0.01F
D1: VISHAY 10MQ100N L1: WURTH ELEKTRONIK 744066330 M1: VISHAY SILICONIX Si7113DN Q1: ZETEX FMMT593 Q2: ZETEX FMMT493 C1, C2: MURATA GRM42-2x7R225
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LT3956 package DescripTion
(Reference LTC DWG # 05-08-1836 Rev C)
28 27 25 24 23 21 20 0.70 0.05
UHE Package Variation: UHE28MA 36-Lead Plastic QFN (5mm 6mm)
30 5.50 0.05 4.10 0.05 31 32 33 34 35 36 3.00
1.88 0.05 0.05 0.12 0.05 3.00
1.53 0.05 0.05
17 16 15 14 PACKAGE OUTLINE 13 12
1.50 REF
0.48
0.05
1
2
3
4
6
0.50 BSC 5.10 6.50 0.05 0.05
8 9 0.25 0.05 2.00 REF
10
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 0.75 5.00 0.10 0.05 PIN 1 NOTCH R = 0.30 OR 0.35 45 CHAMFER 1 1.88 0.10 3.00 0.12 0.10 2 0.10 3 4
R = 0.10 TYP 28 27 2.00 REF 25 24
30
31
32
1.50 REF 33 34 35
36
PIN 1 TOP MARK (NOTE 6)
6.00
0.10
23 0.48 21 20 1.53 0.10 3.00 0.10 0.10
6 8 R = 0.125 TYP 9 10 0.40 0.10 17 16 15 0.25 0.05 0.50 BSC 14 13 12
(UHE28MA) QFN 0110 REV C
0.200 REF 0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 5. EXPOSED PAD SHALL BE SOLDER PLATED 2. DRAWING NOT TO SCALE 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION 3. ALL DIMENSIONS ARE IN MILLIMETERS ON THE TOP AND BOTTOM OF PACKAGE 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LT3956 Typical applicaTion
Buck Mode 1A LED Driver with High Dimming Ratio and Open LED Reporting
VIN 24V TO 80V 100 1M EN/UVLO 61.9k ISN FB VREF 30.1k CTRL 10k INTVCC 100k VMODE SW PWMOUT LT3956 Q1 1k 6 WHITE LEDs 20W M1 VIN ISP 750 0.1 1A Q2 20k 200k 200k 200k C2 4.7F 5 EFFICIENCY (%) 96
Efficiency vs VIN
92
88
84 L1 33H D1 C1 1F 2 VIN 80
20
30
40
50 VIN (V)
60
70
80
0.1F 28.7k 375kHz 47k 2.2nF
PWM SS RT VC GND INTVCC PGND INTVCC 4.7F
3956 TA07a
3956 TA06b
D1: DIODES INC B1100/B L1: WURTH 74456133 M1: VISHAY SILICONIX Si5435BDC Q1: ZETEX FMMT493 Q2: ZETEX FMMT593 C1: TDKC3226X7R2A105K C2: TDKC3225X7RIE475K
relaTeD parTs
PART NUMBER LT3756/LT3756-1/ LT3756-2 LT3755/LT3755-1/ LT3755-2 LT3474 LT3475 LT3476 LT3477 LT3478/LT3478-1 DESCRIPTION 100VIN , 100VOUT, Full Featured LED Controller 40VIN , 75VOUT, Full Featured LED Controller 36V, 1A (ILED), 2MHz, Step-Down LED Driver Dual 1.5A (ILED), 36V, 2MHz Step-Down LED Driver COMMENTS VIN: 6V to 100V, VOUT(MAX) = 100V, True Color PWM Dimming = 3000:1, ISD < 1A, 3mm x 3mm QFN-16 and MS16E Packages VIN: 4.5V to 40V, VOUT(MAX) = 60V, True Color PWM Dimming = 3000:1, ISD < 1A, 3mm x 3mm QFN-16 and MS16E Packages VIN: 4V to 36V, VOUT(MAX) = 13.5V, True Color PWM Dimming = 400:1, ISD < 1A, TSSOP16E Package VIN: 4V to 36V, VOUT(MAX) = 13.5V, True Color PWM Dimming = 3000:1, ISD < 1A, TSSOP20E Package
Quad Output 1.5A, 36V, 2MHz High Current LED Driver VIN: 2.8V to 16V, VOUT(MAX) = 36V, True Color PWM Dimming = 1000:1, with 1000:1 Dimming ISD < 10A, 5mm x 7mm QFN Package 3A, 42V, 3MHz Boost, Buck-Boost, Buck LED Driver 4.5A, 42V, 2.5MHz High Current LED Driver with 3000:1 Dimming VIN: 2.5V to 25V, VOUT(MAX) = 40V, Dimming = Analog/PWM, ISD < 1A, QFN and TSSOP20E Packages VIN: 2.8V to 36V, VOUT(MAX) = 42V, True Color PWM Dimming = 3000:1, ISD < 3A, TSSOP16E Package
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0 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
LT 0510 * PRINTED IN USA
www.linear.com
LINEAR TECHNOLOGY CORPORATION 2010

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